Updated: 25 July 98
to The Project index

09 May 95

   This power amplifier design offers not only extremely low
distortion, but also operation in either Class-A or Class-B. In the
Class-A mode, some new insights have been exploited to allow
unusually linear handling of the situation if the push-pull current
capability is exceeded. The theoretical basis of the design is
explained in detail in the Electronics World article "A TRIMODAL
Power Amplifier" published in two parts in May and June 1995.
   As before, the basic philosophy is the linearisation of each
stage with a high degree of local negative feedback before closing
the global feedback loop; this allows the global feedback factor to
be kept relatively low to maximise HF stability, while still
showing exceptionally good linearity. The main innovative features
are the mode-switching capability, and the unusually low distortion
in Class-AB.

   The PCB follows exactly the design published in Electronics
World for May and June 1995, including the SOAR (safe operating
area) protection against output short-circuits, and the inclusion
of rail fuses. The SOAR protection locus allows output power to be
increased up to approx 40W in 8-Ohms without premature protection
coming in; however be aware that running the design in Class A/AB
mode at this sort of power level is going to need very large
heatsinks for the output devices.
   Please note that it is not in general possible for us to give
advice on modifications to the circuitry.

   The board is of high-quality roller-tinned fibreglass format
with a full silk-screen component ident and solder-mask to minimise
the possibility of solder shorts. Improvements over the previous
Class-B PCB have been made as follows:
  The form-factor of the PCB has been altered so that two will fit
   side by side in a 19" rack chassis.
  All transistor positions have emitter, base and collector marked
   on the top-print to aid fault-finding. TO3 devices are also
   identified on the copper side.
  Wire links have been numbered to make it easier to check they
   have all been fitted.

   The space available for articles in EW is a finite resource, and
so extra information for which no room could be found is given here
about the details of the circuitry and its operation.

   The input stage follows my design methodology in running at a
high tail current to maximise transconductance, and then
linearising it by adding input degeneration resistors R2,3  that
reduce the final transconductance to a suitable level. A current-
mirror TR10,11 forces the collector currents of the two devices to
be equal, thus balancing the input pair and ensuring that it does
not generate second-harmonic distortion; even a small Ic imbalance
generates second-harmonic at a much higher level than the
unavoidable third harmonic. The mirror is degenerated by R6,7 to
minimise the effects of Vbe mismatches in TR10,11; the value of
R6,7 is not critical and any value in the range 10 - 68 Ohms should
be sufficient.
   As a result of the insights gained while studying the slew-rate
characteristics of this configuration, I have increased the input-
stage tail current from 4 to 6 mA, and increased the VAS current
from 6 to 10 mA over the original Class-B circuit. This increases
the maximum slew rate, though as described elsewhere [2] the full
theoretical speed increase is virtually impossible to obtain. One
reason is feedthrough in the VAS current source; in the original
circuit an unexpected slew-rate limit was set by fast edges
coupling through the current source c-b capacitance to reduce the
bias voltage at the worst possible time.
   This effect is minimised in this design by using the negative-
feedback type of current source bias generator, with VAS collector
current as the controlled parameter. TR21 senses the voltage across
R13, and if it attempts to exceed the Vbe, it turns on further to
pull up the bases of TR1 and TR5. C11 filters the rail ripple from
the supply to this circuit and prevents ripple injection from the
V+ rail. 
   The simple act of making the controlled quantity the VAS current
rather than the input tail-current actually contains a beautifully
subtle trap for the unwary. Distortion may be increased in one of
two ways; either "molehill distortion" in which lumps appear on the
distortion residual as the VAS approaches positive clipping, or a
general introduction of second-harmonic. Which effect appears seems
to vary from amplifier example to example and probably depends on
beta spreads. Both effects are due to TR21 altering its collector
voltage to keep the VAS current constant; the VAS operating
conditions vary widely compared with the devices in the input
stage, and the latter has its linearity degraded by the varying
bias voltage. R5, C14 provide decoupling to remove these bias
perturbations and give a complete cure to either manifestation of
the problem. This effect seems too obscure and specialised to add
to the list of Great Distortions, which has now reached eight.

   Note that the input pair positions on the PCB are laid out for
base-in-the-middle TO92 transistor packages. If the high-beta
2SA970 or similar devices are used, they will be found to have base
and collector swopped, so the transistor legs will need to be
sleeved then crossed over.

   There are positions on the PCB for the input bootstrap
components, (C15,R47) but these may simply be omitted if the
facility is not required; for example, a competent preamp with a
buffered output should be able to drive the 2K2 input impedance
without problems. If desired R1,R8,R40 and C2 may be returned to
the original values used in the Class-B amplifier, which gives a
high input impedance without bootstrapping; this will of course
degrade the DC offset and noise performance back to their 
original (but still very respectable) levels.

   The VAS is linearised by beta-enhancing stage TR12, which
increases the amount of local NFB through Miller dominant-pole
capacitor C3. (often referred to as Cdom in these writings) Note
that R36 has been increased to 2K2 to minimise power dissipation,
as there is no significant effect on linearity or slewing. Do not,
however, attempt to omit it altogether, or linearity will be
affected and slewing much compromised. I have tried replacing it
with a constant-current source, but this seems to give no benefits
in linearity.
   As described in [3], the simplest way to prevent ripple from
entering the VAS via the V- rail is old-fashioned RC decoupling,
with a small R and a big C. We have some 200 mV in hand in the
negative direction, compared with the positive, and expending this
on voltage-drop through the RC decoupling will give symmetrical
clipping. R37 and C12 perform this function; the low rail voltages
in this design allow the 1000uF C12 to be a fairly compact
component. 470uF is in general adequate, but the slight increase in
output ripple is measurable. (if not audible)

   The output stage is of the Complementary Feedback Pair (CFP)
type, which gives better linearity and much better quiescent
stability than the more common EF alternative, as a result of the
local negative feedback loops around driver and output device.
Where there is negative feedback there is always the possibility of
oscillation, so this configuration is slightly less straightforward
to apply, which may account for its lack of popularity.
   When oscillation in a CFP stage does occur, it will probably be
at a much higher frequency than global-feedback Nyquist oscillation
(say 2 MHz rather than 400 KHz, but these are VERY rough figures)
and is often very sharply confined to one half-cycle only, due to
the parameter differences that exist even between nominally
complementary devices. I have always found that the 100-Ohm base-
stopper resistors (R12,R24 in the circuit) are a complete
preventative against this sort of misbehaviour, but it is possible
that with some combinations of devices this value may need
   The main function of R25,26 is to define a suitable quiescent
collector current for the drivers TR6,8, and to pull charge
carriers from the output device base when that half of the circuit
is in the process of turning off. The value is not normally
critical, and may be in the region 47 - 100 Ohms, though I have
only tested the circuit for the range 68 - 100. Lower values give
faster output device turn-off, possibly at some expense in basic
linearity; this is of course only relevant to the Class-B mode.
   Several rude things have been said about my output network (ie
the Zobel network C6,R18, and the output inductor L1 with its
damping resistor R19) but I have left it unaltered as while it may
be sub-optimal in some respects, IT WORKS, conferring what appears
to be complete immunity to instability with capacitative loading.
   I suspect that L1 could be reduced in value and size, without
any ill-effects, but testing this against all possible real-life
loads is very lengthy, so there is a powerful incentive to stick
with the output network that worked for your Grandfather...

   SOAR (Safe Operating ARea) protection is given by the networks
around TR18,TR19. This is a conventional single-slope SOAR system,
which effectively draws a straight line protection locus across the
Vce-Ic graph. This is much more effective than simple current-
limiting, but does not follow well the shape of the SOAR area,
which is bounded by two curves, one set by maximum power, and the
other by second-breakdown. Fortunately, in this application, with
its low rail voltages, maximum utilisation of the transistor SOAR
capabilities is not really an issue; the most important thing is to
observe maximum junction temperatures in the A/AB mode. The extra
complexity of dual-slope SOAR limiting was therefore considered
   In the positive half, TR18 monitors the current through TR7 (via
R22) and also the voltage across it. (via R21) D7,8 prevent
spurious conduction of TR18,19 when they are reverse biased
collector-emitter by large output voltage excursions.
   In the negative half of the output stage, TR19 conducts when the
protection locus is exceeded. TR17 is a current limiter for the VAS
transistor TR4; when protection transistor TR19 conducts large
currents can potentially be drawn through TR4. TR17 monitors the
VAS current through R38, and turns on to shunt base drive away from
the VAS when required. This protection is not required for positive
SOAR limiting as TR5 is a current-source with inherently controlled
output current.
   There is actually a subtle trap waiting for those who apply SOAR
limiting incautiously; TR18,19 can conduct just a trifle in normal
operation, and this is usually not very symmetrical so it looks
like second-harmonic distortion. You can drive yourself mad looking
for the source of this distortion in the VAS or the output stage,
and so I strongly recommend that D7,D8 are omitted until the
circuit linearity has been checked.

   It is often stated in hifi magazines that both valve and
semiconductor amplifiers sound better after hours, days, or
possibly even months, of warm-up time. However, regular readers
will have concluded by now, much as I have, that if an unsupported
statement appears repetitively in hifi magazines it is almost
certainly not true, and I fear this is just another example. It
would certainly be possible to come up with a transistor amplifier
with bad crossover distortion and a very poorly-controlled
quiescent current, which might produce audible changes over time,
but the effect would need to be so gross that it would be
impossible to miss in even the most casual measurements, and I have
never seen this reported as an explanation. Such a case would have
to represent plain incompetence in design, for there are a wealth
of electronic techniques for rendering the circuit performance more
or less immune to variations in beta and temperature.
   Since this sort of accusation is usually applied with particular
force to solid-state Class-A designs, because it is clear that the
large heatsinks required take some time to reach their final
temperature, I think it important to state emphatically that in
this design the Class-A electrical operating conditions stabilise
in less than a second, giving the full intended performance. No
"warm-up time" beyond this is required; obviously the heatsinks may
take a long time to reach thermal equilibrium with ambient, but, as
described in the notes above, stern measures have been taken to
ensure that device/heatsink temperature has no effect on operating

   The published design is not intended as a purely cookbook
project- such are not the domain of Electronics World- and
therefore omits the important ancillary of relay muting and DC
offset protection under fault conditions. It is VERY strongly
recommended that a DC protection system is added; eg a ready-
designed version such as the Maplin version, which is known as
Velleman Kit K4700 (Order Code VE24B)

   If you intend to design your own relay protection system- and
this is fairly straightforward- then here are some points to
ponder. Having paid for the relay, for DC protection, it seems
sensible to use it for system muting as well, to prevent thuds and
bangs from the upstream parts of the audio system from reaching the
speakers at power-up and power down. The amplifier itself, being
dual-rail (ie DC-coupled) does not generate large thumps itself,
but it cannot be guaranteed to be completely silent.
   Your relay-control system should:

1 Leave the relay de-energised when muted. At power-up, there
should be a delay of at least 1 second before the relay closes.
This can be increased if required.

2 Drop out the relay as fast as is possible at power-down, to stop
the dying moans of the preamp etc from reaching the outside world.
My own preference is to do this by sensing the incoming AC; when
this disappears, the relay is dropped out within 10 msec, which
should be long before the various reservoir capacitors in the
system can begin to discharge. However, if the mains switch
contacts are generating RF that is in turn reproduced as a click by
the preamp, then this system may not be fast enough to mute it.

3 Drop out the relay as fast as is possible when a DC offset of
more than 1 - 2 V, in either direction, is detected at the output
of either power amp channel; the exact threshold is not critical.
This is normally done by low-pass filtering the output (47K and 47
uF works OK) and applying it to some sort of absolute-value circuit
to detect offsets in either direction. The resulting signal is then
OR-ed in some way with the muting signal mentioned above.

4 Don't forget that the contacts of a relay have a much lower
current rating for breaking DC rather than AC. This is an issue
that doesn't seem to have attracted the attention it deserves.

   Please note that the HT rail fuses are intended only to minimise
amplifier damage in the event of output device failure. They must
not be relied upon for speaker protection against DC offset faults.

   Fuses in series with the output line are sometimes recommended
for DC offset protection. It appears to be true that they have a
better chance of saving expensive loudspeakers than the HT fuses,
but there are at least three snags:

1 Selection of the correct fuse size is not at all easy. If the
fuse rating is small and fast enough to provide some real
protection, then it is likely to be liable to nuisance blowing on
large bass transients. A good visual warning is given by behaviour
of the fuse wire; if this can be seen sagging on transients, then
it is going to fail sooner rather than later. At least one writer
on Class-A amplifiers gave up on the problem, and coolly left the
tricky business of fuse selection to the constructor!

2 It has been widely reported that fuses running within sight of
their rated current generate distortion at LF due to changes in
resistance caused by RMS heating; this is presumably third
harmonic. It is of course true that many widely reported things in
the audio field have no more existence than the unicorn, but the
mechanism does (for once) sound plausible, so although I have no
data on this to offer myself, I think it desirable to make the
point. This problem can, in theory at least, be sidestepped by
putting the fuse inside the global feedback loop; however what will
the amplifier do when its global feedback is abruptly removed when
the fuse blows?

3 Fuses obviously have significant resistance (otherwise they
wouldn't blow) so putting one in series with the output will
degrade the theoretical damping factor. However, whether this is of
any audible significance is very doubtful.

   There is naturally a great deal more that could be said about
amplifier protection, and if I am spared I hope to deal with some
of the lesser-known issues in a future article.

   The amplifier design as published yields 20W rms into 8 Ohms,
but may be configured for other powers by appropriate choice of
supply rail voltage and quiescent current. I gave an upper limit of
30W in the first part of the article, but on mature reflection this
seems a bit high for one output pair, and it might be best to
reduce the quiescent current somewhat from the 6-Ohm case.
   The choice of the appropriate quiescent current for a given
output voltage capability depends on an appreciation of the load
impedances that are likely to be encountered in real life; the
published design assumes that a current capability that allows 6-
Ohm resistive loads to be fully driven is adequate. Remember that
unlike some Class-A amplifiers, this one does not run abruptly into
horribly audible current-clipping when it runs out of push-pull Iq
into a low-Z load, but goes instead into a low-distortion Class-AB.
   In general, if the supply rail voltages are increased by, say
10%, then the Class-A/AB mode quiescent current must be similarly
increased by 10% to maintain the same load-driving capability. Note
that the quiescent dissipation in this mode has now increased by
approx 20%, as we have upped both voltage and current, and this
needs to be taken into account when arranging the heatsinking.

   This design has excellent supply-rail rejection, and so a simple
unregulated supply is perfectly adequate. The use of regulated
supplies is definitely unnecessary, and I would recommend strongly
against their use. At best, you have doubled the amount of high-
power circuitry to be bought, built, and tested. At worst, you
could have intractable HF stability problems, peculiar slew-
limiting, and some expensive device failures. Just say no!

   If you wish to modify the unregulated PSU design given, then
remember that in the Class-A/AB mode, a heavy current is drawn
continuously; rectifier power dissipation and reservoir capacitor
ripple-current capability must be taken much more seriously than is
usual for Class-B. In general, 10,000uF is an absolute minimum for
the reservoirs, and 20,00uF is strongly recommended. Increases
beyond this will do no harm (so long as the turn-on surge can be
handled by the rectifiers) and will marginally increase the
unclipped output power.

   When selecting the value of the mains fuse in the transformer
primary circuit, remember that toroidal transformers take a large
current surge at switch-on. The fuse will definitely need to be of
the slow-blow type.

   As before, we have attempted to configure the PCB to use easily
obtainable components, in particular the following, which are all
available from Maplin Electronics. Maplin order codes are given for

1) Driver heatsinks. The PCB has mounting holes suitable for
   heatsink Type-SW38-2                       (Order Code JW28F)
2) Fuseholder clips. 20mm Fuse Clip Type 1,   (Order Code WH49D)
3) Quiescent-adjust preset. Cermet preset 1K, (Order Code WR40T)
4) Wirewound resistors. 3W "WW Min"           (Order Code W+value)
5) Non-electrolytic capacitors; Polyester.    (eg Order Code WW41U)
6) Output inductor; 18 swg enamelled copper wire.(Order code BL25C)

   The mechanical design adopted depends very much on personal
taste and resources, but I will offer a few points that need to be
taken into account:

1 A Class-A amplifier requires extensive heatsinking with a free
convective air flow, and this points toward putting the sinks on
the side of the amplifier; the front will carry at least the mains
switch and power indicator light, while the back carries the in/out
and mains connectors, so only the sides are completely free.
   The internal space in the enclosure will require some
ventilation to prevent heat buildup; slots or small holes are
desirable to keep foreign bodies out. Avoid openings on the top
surface as these will allow the entry of spilled liquids, and
increase dust entry. BS415 is a good starting point for this sort
of safety consideration, and this specifies that slots should be
not more than 3mm wide.

2 A toroidal transformer is strongly recommended because of its
low external field. It must be mounted so that it can be rotated to
minimise the effect of what stray fields it does emit. Most
suitable toroids have single-strand secondary lead-outs, which are
too stiff to allow rotation; these can be cut short and connected
to suitably-large flexible wire such as 32/02, with carefully
sleeved and insulated joints. One of our prototypes had an ILP
toroid mounted immediately adjacent to the TO3 end of the amplifier
PCB; however complete cancellation of magnetic hum (output level
below -90 dBu) was possible on rotation of the transformer.
   A more difficult problem is magnetic radiation caused by the
reservoir charging pulses (as opposed to the ordinary magnetisation
of the core, which would be essentially the same if the load
current was sinusoidal) which can be picked up by either the output
connections or cabling to the power transistors if these are
mounted off-board. For this reason the transformer should be kept
physically as far away as possible from even the high-current
section of the amplifier PCB.
   As usual with these transformers, make certain that the bolt
through the middle cannot form a shorted turn by contacting the
chassis in two places.

   The distortion performance of a Class-B or AB power amplifier
depends almost as much on the topology and layout of the power and
ground wiring as on the subtleties of the circuit design. (Things
are much simpler if you can assume that the amplifier will never
leave Class-A, but with realistic quiescent currents this is not a
safe assumption) This has been taken into account in the PCB
layout, but the external wiring has to be the responsibility of the
constructor. We therefore give a recommended wiring scheme that has
been approved by the designer. (The assumption is made that a
simple unregulated supply is used; as noted above, a regulated
supply is quite unnecessary and may cause unforeseen complications)

1 There are several important points about the wiring for any
power amplifier; see the attached wiring diagram:
 a: Keep the + and - supply wires to the amplifiers close together.
This minimises the generation of distorted magnetic fields which
may otherwise couple into the signal wiring and degrade linearity.
Sometimes it seems more effective to include the 0V line in this
cable run; if so it should be tightly braided to keep the wires in
close proximity.
   For the same reason, if the power transistors are mounted off
the PCB, the cabling to each device should be configured to
minimise loop formation. 

 b: The rectifier connections should go direct to the reservoir
capacitor terminals, and then away again to the amplifiers. Common
impedance in these connections superimposes charging pulses on the
rail ripple waveform, which may degrade amplifier PSRR.
 c: Do not use the connection between the two reservoir capacitors
as any form of star point. It carries heavy capacitor-charging
pulses that generate a significant voltage drop even if thick wire
is used. As the drawing shows, the "star-point" is tee-ed off from
this connection. This is a star-point only insofar as the amplifier
ground connections split off from here, so do not connect the input
grounds to it, as distortion performance will suffer.

2 Driver transistor installation. These should be mounted onto
their separate heatsinks with silicone thermal washers, to ensure
good thermal contact. Use the spring clips intended to hold the
package firmly against the sink. Electrical isolation between
device and heatsink is not essential, as the PCB makes no
connection to the heatsink fixing pads, but you will get it anyway
unless the washer is damaged.

3 TO3 power transistor installation. The PCB layout allows the
TO3s to be mounted on an aluminium thermal-coupling flange which is
bolted to the PCB, and the TO3 pins then soldered directly in.
Alternatively, the TO3s can be mounted off-PCB (eg if you already
have a large heatsink with TO3 drillings) with wires taken from the
TO3 pads on the PCB to the remote devices. These wires should be
fastened together (two bunches of three is fine) to prevent loop
formation; see above. I cannot give a maximum safe length for such
cabling; certainly 8 inches causes no stability problems. The
emitter and collector wires should be substantial, such as 32/02,
but the base connections can be as thin as 7/02 without problems.

   It is recommended that the flange is drilled with suitable holes
to allow bolts to pass through the TO3 fixing holes, through the
flange, and then be secured by nuts and crinkly washers which will
ensure good contact with the PCB mounting pads. Insulating sleeves
are essential around these bolts where they pass through the
flange; nylon is a good material for these as it has a good high-
temperature capability. Depending on the size of the holes drilled
for the two TO3 package pins, (and this should be as small as
practicable to maximise the area for heat transfer) these are also
likely to require insulation; silicone rubber sleeving carefully
cut to length is very suitable.
   An insulating thermal washer must be used between TO3 and
flange; these tend to be delicate and the bolts must not be over-
tightened. If you have a torque-wrench, then 10 Newton/metre is an
appropriate upper limit for M3.5 fixing bolts. Do not solder the
two transistor pins to the PCB until the TO3 is firmly and
correctly mounted, fully bolted down, and checked for electrical
isolation from the heatsink. Soldering these pins and then
tightening the fixing bolts is likely to force the pads from the
PCB. If this should happen then it is quite in order to repair the
relevant track or pad with a small length of stranded wire to the
pin; 7/02 size is suitable for a very short run.

4 Bias-generator transistor TR13 mounting. Our previous design
(The Class-B amplifier; PCB-001) used a double emitter-follower or
EF output stage; for this the optimal place to mount sensor TR13
for effective thermal compensation was the top of the TO3 cans, to
get as close as possible to the output device junction temperature.
This is mechanically awkward, and not necessary here.
   The TRIMODAL design uses a CFP output stage rather than EF, to
increase both output efficiency and linearity. The output device
junction temperature is now almost irrelevant, being servo-ed out
by the local CFP feedback loop, and in Class-B mode the
temperature-sensor TR13 must now aspire to reach the temperature of
the drivers instead, which is mechanically much simpler. A position
for mounting TR13 on the other side of the TR6 heatsink (HS1) is
provided on the PCB, and a second thermal washer and spring clip
are required for mounting.
   If this method of mounting is used, then obviously there is a
thermal delay and attenuation between the driver and the sensor,
due to the thermal mass and convective losses of the heatsink
coming between the two devices. A better solution for optimal
thermal compensation is to mount the sensor on top of the driver
transistor package, ie on the same side of the heatsink. Flying
leads are then run back to the original TR13 position. The standard
spring clip has enough give (just) to allow the extra transistor
and extra thermal washer to be slipped between it and the driver
package. The implications of this improvement will hopefully be
further explored in a future article.
   In Class-A/AB mode the quiescent current is controlled by
another negative-feedback loop, which can measure the current it is
controlling directly, and so this sort of temperature compensation
is not an issue.

   NOTE: Make sure TR13 is properly in contact with the surface it
is sensing. Without thermal compensation the quiescent stability in
Class-B will be seriously degraded, though almost certainly not to
the point where thermal runaway is possible.

   The amplifier design presented here is inherently safe in that
all the DC voltages are too low to present any kind of electric-
shock hazard. However, there are a few points I think the
constructor should consider.

1 The supply rails are low-voltage, but the reservoir capacitors
store a significant amount of energy. If they were to be shorted
out by a metal finger-ring then a nasty burn is likely. If your
bodily adornment tends toward the metallic then it should be
removed before diving into the amplifier.

2 Any amplifier containing a mains power supply is potentially
lethal. The amplifier is unusual and slightly more complex than
some, so the risks involved in working for some time on the
powered-up chassis must be considered. The metal chassis MUST be
securely earthed to prevent it becoming live if a mains connection
falls off, but this presents the snag that if one of your hands
touches live, there is a good chance that the other is touching
chassis ground, so your well-insulated training shoes will not save
you. All mains connections (neutral as well as live) must therefore
be properly insulated so they cannot be accidentally touched by
finger or screwdriver. My own preference is for double insulation;
for example, the mains inlet connector not only has its terminals sleeved, but there is also an overall plastic boot fitted over the
rear of the connector, and secured with a ty-wrap.
   Note that this is a more severe requirement than BS415 which
only requires that mains should be inaccessible until you remove
the cover.(With a tool, though a coin is permissible)

3 A Class-A amplifier runs HOT and the heatsinks may well rise
above 70 degC. This is not likely to cause serious burns, but it is
painful to touch. You might consider this point when arranging the
mechanical design.

4 Note the comments on slots and louvres in the section on
"Mechanical Design" above.

5 Readers of hifi magazines are frequently advised to leave
amplifiers permanently powered for optimal performance. Unless your
equipment is afflicted with truly doubtful control over its own
internal workings, this is quite unnecessary. While there should be
no real safety risk in leaving a soundly-constructed power
amplifier powered permanently, I see no point and some risk in
leaving unattended equipment powered; in Class-A/AB mode, there may
of course be an impact on your electricity bill...

1 By far the most important step to successful operation is a
careful visual inspection before switch-on. As in all power
amplifier designs, a wrongly-installed component may easily cause
the immediate failure of several others, making fault-finding
difficult, and the whole experience generally less than
satisfactory. It is therefore most advisable to meticulously check:
  That the supply and ground wiring is correct.
  That all transistors are installed in the correct positions.
  That the drivers and TO3 output devices are not shorted to their
respective heatsinks through faulty insulating washers.
  That the circuitry around the bias generator TR13 in particular
is correctly built. An error here that leaves TR13 turned off will
cause large currents to flow through the output devices and may
damage them before the rail fuses can act.
   I recommend that the initial testing is done in Class-B mode.
There is the minimum amount of circuitry to debug (The Class-A
current-controller can be left disconnected, or not built at all
until later) and at the same time the Class-B bias generator can be
checked for its operation as a safety-circuit on Class-A/AB mode.

2 The second stage is to obtain a good sinewave output with no
load connected. A fault may cause the output to sit hard up against
either rail; this should not in itself cause any damage to
components. Since a power amp consists of one big feedback loop,
localising a problem can be difficult. The best approach is to take
a copy of the circuit diagram and mark on it the DC voltage present
at every major point. It should then be straightforward to find the
place where two voltages fail to agree; eg a transistor installed
backwards usually turns fully on, so the feedback loop will try to
correct the output voltage by removing all drive from the base. The
clash between "full-on" and "no base-drive" signals the error.
   When checking voltages in circuit, bear in mind that C2 is
protected against reverse voltage in both directions by diodes
which will conduct if the amplifier saturates in either direction.
   This DC-based approach can fail if the amplifier is subject to
high-frequency oscillation, as this tends to cause apparently
anomalous DC voltages. In this situation the use of an oscilloscope
is really essential. An expensive oscilloscope is not necessary; a
digital scope is at a disadvantage here, because HF oscillation is
likely to be aliased into nonsense and be hard to interpret.

3 The third step is to obtain a good sinewave into a suitable
high-wattage load resistor. It is possible for faults to become
evident under load that are not shown up in Step 2 above.
   Setting the quiescent current for any Class-B amplifier can only
be done accurately by using a distortion analyser. If you do not
have access to one, the best compromise is to set the quiescent
voltage-drop across both emitter resistors (R16,17) to 10mV when
the amplifier is at working temperature; disconnect the output load
to prevent DC offsets causing misleading current flow. This should
be close to the correct value, and the inherent distortion of this
design is so low that minor deviations are not likely to be very
significant. This implies a quiescent current of approx 50 mA.

4 When you are satisfied that the amplifier is working correctly
in Class-B, the Class-A/AB mode can be tested. The voltage across
both output emitter resistors R16,17 should now be increased to
300mV (the same as the voltage across R31) and it should remain
very nearly constant as the amplifier warms up. 

5 It may simplify faultfinding if D7,D8 are not installed until
the basic amplifier is working correctly, as errors in the SOAR
protection cannot then confuse the issue. This demands some care in
testing, as there is no short-circuit protection. 

   Two mounting holes for fixing the PCB to the chassis are
provided on the input edge. These accept standard plastic pillars.
Four further fixing holes are provided for fixing the PCB to the
heatsink; if the power transistors are mounted off-board then the
outer two of these can be used for two more mounting pillars.

   Since the component selection, construction, and usage of this
PCB are entirely outside our control, we can accept no
responsibility for the functioning or performance of amplifiers
constructed with it. We are therefore unable to enter into
correspondence regarding faultfinding, substitute components, etc.
   We can accept no liability for loss, damage or injury incurred
by the construction or operation of this design.

ADDENDUM TO TRIMODAL AMPLIFIER NOTES. 19 JUNE 95 PCB002A1.DOC Some minor corrections and the fruits of experience... 1) Final testing showed that R22 was not required. It was therefore omitted from the published circuit, though there is still a position for it on the PCB. Fit a wire link in this position. 2) Owing to a production oversight, there are three pads (on links LK9,10) which are covered in solder resist. These should be scraped clean with a sharp knife. Do it just before soldering so the copper does not have time to oxidise. 3) If high-beta input transistors with the ECB pinout are used, it will be necessary to cross the C and B legs before fitting them in the PCB. The obvious insulation to use is 1mm bore silicone sleeving- but actually this is not a good idea. The square-section transistor legs seem to be able to cut through this with surprising ease, and I have found there is every chance of installing two short-circuits along with the components. Two simultaneous faults make faultfinding difficult, so be warned... 4) A Test Mode jumper position (J1) has been added next to the preset; this has the correct pitch for standard pins and push-on jumpers. The jumper must be removed once you are happy the Class-A current controller is working, and the preset turned back to the appropriate position for Class-B biasing. If you don't remove it, TR13 is still active as a Vbe-multiplier in Class-A mode, and it will be impossible for enough quiescent current to flow. 5) There is a minor error in Part 2 of the article. On the second page the use of a 2.56V band-gap reference is discussed; please note that R30 must be increased to 7K (not 5K as stated) in order to get the desired 300mV across R31. 6) Detailed evaluation of the PCB has shown that the grounding of C12 could be improved. If the track to the +ve terminal is cut, and a wire run back from here directly to the 0V supply pad, the -ve HT ripple rejection is improved by several dB. Not essential, but recommended for the perfectionist. 7) As with all Class B or AB amplifiers, layout of supply wiring is critical if the best distortion performance is to be obtained. In PCB tests, the best results were got by taking the +ve HT wire directly off to the supply, while the -ve HT wire was run up the PCB parallel with R16,17 (NB these get hot) and fastened it to the +ve wire at the top PCB edge. The 0V cabling run appeared to be much less critical, and was put into the same fastening; the 3 wires then go to the PSU reservoirs in a tight bundle. Douglas Self.